Isolation and signal filter transformer

ABSTRACT

One embodiment of the invention includes a transformer that is at least partially formed on a substrate. The transformer is for electrical isolation and signal filtering. The transformer includes the following elements. A portion of the substrate has two holes through the substrate. A ferrite core is formed in a loop through the two holes. A primary winding is made of a first metal trace. The first metal trace is formed in a spiral around one hole. The first metal trace has at least two primary terminals acting as an input to the transformer. The transformer also includes a secondary winding. The secondary winding is made of a second metal trace. The second metal trace is formed in a spiral around the other hole. The second metal trace has at least two secondary terminals acting as an output from the transformer. The parasitic capacitance of the first metal trace and the magnetizing inductance of the primary winding cause signals received at the primary terminals to be filtered as the signals are passed through the transformer to the secondary terminals.

RELATED APPLICATIONS

This invention is a continuation - in- part of U.S. patent application Ser. No. 08/641,375 now U.S. Pat. No. 5,801,602, entitled "Isolation and Signal Filter Transformer," filed Apr. 30, 1996.

THE FIELD OF THE INVENTION

This invention relates to the field of communications circuits. In particular, the invention relates to a use of an isolation and filter circuit.

BACKGROUND OF THE INVENTION

Isolation circuits and filter circuits are often used in devices connected to transmission lines. The transmission lines carry a signal to the devices or away from the devices. In a network adapter card, for example, a combination isolation and filter circuit is used to transfer and filter a signal from the network cable to the adapter card, or from the adapter card to the network cable. The isolation part of the circuit isolates the rest of the adapter card circuitry from the network cable. The filter part removes high frequency components of the signal.

FIG. 1 illustrates two combination isolation and filter circuits 100 as used in a 10 MHz Ethernet adapter card. The transmit circuit 101 transmits signals from the adapter card to the network cable, while the receive circuit 102 receives signals from the network cable and transmits them to the adapter card. Part number FL1012 and FL1066, available from Valor, Inc., implements the isolation and filter circuits 100.

The transmit circuit 101 includes a filter 110, a transformer 120 and a common mode choke circuit 130. The filter 110 connects to the transmit side of the adapter card electronics and to the transformer 120. The transformer 120 connects to the common mode choke circuit 130. The common mode choke 130 connects to the network cable.

The filter 110 includes number of inductors and capacitor. The inductors and capacitors act as a multi-pole filter for signals being transmitted to the network. The frequency response of the transmit circuit 101 is shown in FIGS. 2A and 2B. The 100 MHz frequency response graph 210 shows that the transmit circuit 101 acts as a low pass filter, with a response that quickly rolls off near 16 MHz. The 1 GHz frequency response graph 200 shows that the transmit circuit 101 does not transmit more than -20 dB for any other frequency between 25 MHz and 1 GHz. For 10 Mbit Ethernet communications, such a frequency response is desirable because high frequency noise components of the transmitted signal are removed before the signal is transmitted to the network cable. The frequency response thus allows the adapter card to meet electromagnetic interference requirements.

Returning to FIG. 1, the transformer 120 electrically isolates the adapter card from the network cable. The transformer 120 has a relatively good frequency response. That is, both high frequencies (e.g., 800 MHz) and low frequencies (20 kHz) are passed through the transformer 120 with little attenuation.

The common mode choke 130 is part of the filter 110 and helps to remove high frequency components from the signal being sent to the network cable.

The receive circuit 102 includes circuits similar to those in the transmit circuit 101. However, the receive filter 112 has fewer inductors and capacitors than the filter 110, resulting in a low pass filter with fewer poles. Having fewer poles means that the receive filter 112 does not have as steep a roll off as the filter 110. Because the receive characteristics are not as stringent as the transmit characteristics, a filter having fewer poles is acceptable. Importantly, reducing the number of inductors and capacitors also reduces the cost of the filter 110.

In high volume manufacturing, saving only a few pennies per adapter card can save millions of dollars per year. The use of the isolation and filter circuit 100 adds a significant cost to the price of the adapter cards. Thus, it is desirable to have an isolation and filter circuit 100 with a lower cost but still maintain a similar frequency response.

Additionally, for different communications standards, a different part is needed to implement the different isolation and filter circuits 100. For example, one part will have the desired frequency response for 10 Mbit Ethernet, while another part will have the desired frequency response for 16 MHz token ring. Maintaining an inventory of all these different parts is expensive because, the individual cost of each of the parts is relatively expensive. Therefore, it is desirable to have a more versatile isolation and filter circuit 100.

A SUMMARY OF THE INVENTION

An isolation and filter transformer is described.

One embodiment of the invention includes a transformer that is at least partially formed on, or in, a substrate. The transformer is for electrical isolation and signal filtering. The transformer includes the following elements. A portion of the substrate has two holes through the substrate. A ferrite core is formed in a loop through the two holes. A primary winding is made of a first metal trace. The first metal trace is formed in a spiral around one hole. The first metal trace has at least two primary terminals acting as an input to the transformer. The transformer also includes a secondary winding. The secondary winding is made of a second metal trace. The second metal trace is formed in a spiral around the other hole. The second metal trace has at least two secondary terminals acting as an output from the transformer. The parasitic capacitance of the first metal trace, the leakage inductance and the magnetizing inductance of the primary winding and the secondary winding cause signals received at the primary terminals to be filtered as the signals are passed through the transformer to the secondary terminals. Thus, one embodiment of the invention greatly reduces the cost of manufacturing filtering and isolation circuits in network adapter cards, and other communications systems, by integrating the filtering and isolation functions into one structure.

In one embodiment, reactive elements, such as an external capacitor, are added to further change the frequency response of the transformer.

In one embodiment, the isolation and signal filter transformer is tuned to have a desired frequency response by taking advantage of the parasitic capacitances and inductances of the traces of both the primary and the secondary winding. The parasitic capacitances and inductances are controlled by the layout of these traces.

Although many details have been included in the description and the figures, the invention is defined by the scope of the claims. Only limitations found in those claims apply to the invention.

A BRIEF DESCRIPTION OF THE DRAWINGS

The figures illustrate the invention by way of example, and not limitation. Like references indicate similar elements.

FIG. 1 illustrates a prior art filter circuit.

FIG. 2a and FIG. 2b illustrate the frequency response of the prior art filter of FIG. 1.

FIG. 3 illustrates one embodiment of the invention as used in a network adapter card.

FIG. 4 illustrates a layout of a transformer.

FIG. 5a and FIG. 5b illustrates a linear model of the layout of FIG. 4.

FIG. 6 illustrates an example frequency response of one embodiment of the invention.

FIG. 7a and FIG. 7b illustrate an example frequency response of one embodiment of the invention including an external capacitor.

FIG. 8a and FIG. 8b illustrate an example frequency response of one embodiment of the invention including two external capacitors.

FIG. 9 illustrates a bifilar transformer.

FIG. 10 illustrates a multi-layered substrate having an integrated transformer.

FIG. 11 illustrates the traces for an integrated filter transformer.

FIG. 12 illustrates one embodiment of an external circuit that can be used with the integrated transformer of FIG. 11.

FIG. 13 illustrates the reflected signal transmission for the circuit of FIG. 12.

FIG. 14 through FIG. 35 describe alternative embodiments of the integrated transformers.

THE DESCRIPTION

Isolation and Filter Circuit

FIG. 3 illustrates a network adapter card using one embodiment of the invention. FIG. 3 includes a computer system 300, with a network adapter card 310, and a network cable 370. The network adapter card 310 includes two integrated transformers that cost much less than previous combined filter and isolation circuits. In one embodiment, the two integrated transformers have their windings formed directly on, and/or in, the network adapter card's substrate. A ferrite, or other metal, core is then positioned through windings, and therefore through the substrate. The parasitic characteristics, such as capacitance, inductance, and resistance, of these windings and the core are used to create a filter.

Before discussing the details of the integrated transformers, a use of these transformers in the computer system 300 is described. The computer system 300 includes a processor 380, a memory 390, a bus 399, and the network adapter card 310. The processor 380, the memory 390 and the network adapter card 310 are all connected to, and communicate over, the bus 399. In one embodiment, the computer system 300 includes one of an IBM PC-compatible computer, a Macintosh™ computer, a Sun SPARCstation™ computer, or some other computer having a processor and a memory.

The network cable 370 includes at least one pair, and in the example of FIG. 3, two pairs, of transmission lines. The TX (transmit) transmission lines 371 communicate signals from the network adapter card 310 to the rest of the network. The RX (receive) transmission lines 372 communicate signals from the rest of the network to the network adapter card 310. In one embodiment of the invention, the network cable 370 includes a 10base-T cable. In another embodiment, the network cable 370 includes one of: an AUI cable, twisted pair FDDI cable, 100base-T cable, and AppleTalk™ cable, or some other type of transmission line medium (e.g., a telephone line).

The network adapter card 310 includes a substrate 360, the TX integrated transformer 321, the RX (receive) integrated transformer 322, and the other adapter electronics 330. The TX integrated transformer 321, the RX integrated transformer 322 and the other adapter electronics 330 are mounted on, in and/or through the substrate 360. Both of the TX integrated transformer 321 and the RX integrated transformer 322 connect to the other adapter electronics 330. The TX integrated transformer 321 also connects to the TX transmission lines 371. The RX integrated transformer 322 connects to the RX transmission lines 372.

In one embodiment, the substrate 360 includes a double-sided printed circuit board made of 0.062+/-0.005 inch thick glass epoxy, natural color, laminated NEMA GRADE FR-4 available from Solectron, Inc. of Milpitas, Calif. The substrate 360 has a dielectric constant of 4.5+/-0.4 at 1 MHz, the dielectric strength perpendicular to an adjacent layer has a minimum of 750 VDC per 0.001 inch thickness of the substrate. Traces are formed on, or in the substrate, with 1/2 ounce copper. In other embodiments, other materials are used, such as, a printed circuit board having more than two layers of copper, a substrate including a ground plane layer, a 0.075+/-0.003 thick glass epoxy, or polyamide fiberglass GRADE FR-5. Other embodiments also have variations in the dielectric constant and the dielectric strength.

The other adapter electronics 330 include the electronics for preparing, sending and/or receiving data over the network cable 370 or the bus 399, and interfacing with the bus 399. In one embodiment, the other adapter electronics 330 includes the circuits for implementing the 10base-T standard, except for those parts of the standard implemented by the integrated transformers. In other embodiment, the other adapter electronics 330 implement other communications standards.

In one embodiment, the computer system 300 does not include the network adapter card 310. In this embodiment, the components on the network adapter card 310 are included directly on the motherboard to which the processor 380 and the memory 390 are mounted.

Integrated Transformer

As mentioned previously, in one embodiment, the integrated transformers are manufactured by forming the windings directly on the substrate and inserting a core through the windings. This process results in a less than ideal transformer. Significant parasitic capacitance, inductance, and resistance are all a result of manufacturing the integrated transformers in this way. However, one embodiment of the invention uses these parasitic characteristics advantageously to create a filter. The filter passes only those parts of a signal that are desired. Thus, the integrated transformers completely, or at least partially, replace the prior art combined isolation and filter circuit 100. This significantly reduces the cost of the network adapter card 310.

FIG. 4 shows a layout of an integrated transformer 400. In other embodiments of the invention, variations of the integrated transformer 400 are used as the TX integrated transformer 321 and the RX integrated transformer 322.

The integrated transformer 400 includes two sets of windings, two holes and a ferrite core 450. The two sets of windings include the primary winding 410 and the secondary winding 420. Note that in another embodiment, the primary winding 410 is used as a secondary winding, while the secondary winding 420 is used as a primary winding. The primary winding 410 is made of a tetrace 412 and a bottom trace 414 that are connected through the substrate 360 by the via 411. Similarly, the secondary winding 420 is made of a top trace 422 and a bottom trace 424 that are connected through the substrate 360 by the via 421.

FIG. 4 presents two top views of a portion of the substrate 360 to show the two traces. The first top view shows the top traces formed as spirals, around the holes, on the top surface of the substrate 360. The bottom traces are formed as spirals, around the holes, on the bottom surface of the substrate 360. In this embodiment, the bottom traces are formed directly under the top traces. (The second top view shows the bottom traces as one would see the bottom traces if the substrate 360 were transparent and the top traces were removed.)

In another embodiment, the bottom traces are not directly under the traces. This changes the parasitic characteristics of the integrated transformer (e.g., as described below, Cp1 570 and L11 520 for the primary winding 410, and Cp2 575 and L12 525 for the secondary winding 420).

Importantly, in one embodiment, the traces are made of the same material as is used in the rest of the network adapter card 310. For example, the top trace 412, the top trace 422, the bottom trace 414, and the bottom trace 424 are made of copper and are formed on the surface of the substrate 360. The traces are made during the manufacturing of the network adapter card 310 and at the same time that traces are made for the other adapter electronics 330. In one embodiment, the top and bottom traces are 0.007+/-0.001 inch wide, having a minimum spacing of 0.008 inches.

Two holes are made in the substrate 360. Each hole is made at approximately the center of a winding. For example, hole 415 is made in the center of the primary winding 410, while hole 425 is made in the center of the secondary winding 420. In the example of FIG. 4, each hole is approximately square. However, in one embodiment, the holes are approximately round because round holes are less costly to manufacture.

A ferrite core 450 is placed in the holes to form a closed loop. In one embodiment, the ferrite core 450 is made of two pieces of ferrite. The top is a U-shaped ferrite core piece 452. The bottom is an I-shaped ferrite core piece 454. The ends of the U-shaped piece 452 are placed through the holes. The I-shaped piece 454 is then attached to the ends of the U-shaped piece 452 that are protruding through the bottom of the substrate 360. In one embodiment, the I-shaped piece 454 is attached to the U-shaped piece using a clamp. The clamped U-shaped piece 452 and the I-shaped piece 454 form a closed ferrite loop through the primary winding 410 and the secondary winding 420. The closed loop allows the magnetic flux generated from the primary winding 410 to flow through the ferrite core 450. As the magnetic flux passes through the secondary winding 420, a current is induced in the secondary winding 420. Thus, a transformer is formed.

In one embodiment, the ferrite core 450 is made of two U-shaped core pieces from Fair-Rite Products Corporation of Wailkill, N.Y., part number 90-002002, made of material 77 (manganese-zinc material).

In this embodiment, the distance between the inner edges of the two holes is slightly less (e.g., 0.005 inches) than 0.09 inches and the distance between the outer edges of the two holes is slightly greater (e.g., 0.005 inches) than 0.34 inches. The traces are 0.007 inches wide. The spacing between two adjacent portions of trace material is 0.008 inches. The distance from the edge of a hole to the nearest edge of a trace is approximately 0.01 inches. Other embodiments of the invention use different sizes and distances depending upon the desired frequency response of the integrated transformer.

Thus, FIG. 4 and the above description, show how an integrated transformer can be built from a few traces, a substrate 360 and a ferrite core 450. This integrated transformer is much less expensive than previous isolation and filter circuits, such as the isolation and filter circuit 100. The reduced costs are a result of having fewer, lower cost, and more reliable components. For example:

the additional capacitors and inductors of the circuit 100 are not needed to provide the desired signal filtering.

the transformer 120 costs significantly more to build than the integrated transformer because the transformer 120 uses relatively expensive wire windings.

the integrated transformer is more reliable than the circuit 100 because the integrated transformer requires few, or no, external capacitors and inductors.

the integrated transformer is built using more reliable technology (e.g., the traces may have better tolerances than the tolerances of the transformer's 120 windings).

because the tolerances for fabricating the substrate 360 tend to be better than for the transformer 120, the integrated transformer has more consistent performance.

Integrated Transformer Model

FIG. 5a and FIG. 5b illustrate an integrated transformer linear model 500. The integrated transformer linear model 500 shows how parasitic characteristics are used to change the filtering characteristics of an integrated transformer.

Table 1 shows the devices of the integrated transformer linear model 500.

                                      TABLE 1                                      __________________________________________________________________________     Label   Description                                                            __________________________________________________________________________     Rs1 510 Series resistance of the entire primary winding 410 (top trace                 412                                                                            resistance plus bottom trace 414 resistance).                          L11 520 Leakage inductance of the primary winding 410.                         Lm1 550 Magnetizing inductance of the primary winding 410.                     Cp1 570 Inter-trace capacitance of the primary winding 410 (as shown in                Figure 5b and also between the top trace 412 and the bottom                    trace                                                                          414).                                                                  Rs2 515 Series resistance of the entire secondary winding 420 (top trace               422                                                                            resistance plus bottom trace 424 resistance).                          L12 525 Leakage inductance of the secondary winding 420.                       Cp2 575 Inter-trace capacitance of the secondary winding 420 (as shown                 in                                                                             Figure 5b and also as between the top trace 422 and the bottom                 trace                                                                          424).                                                                  Z2 501  Impedance of the load connected to the output of the secondary                 winding 420.                                                           Ideal   An ideal transformer with a transformer ratio of 1:1. (Although                other                                                                  Transformer 503                                                                        ratios are used in other embodiments where the output voltage                  needs                                                                          to be stepped up or stepped down.)                                     Cc 505  Capacitive coupling between the primary winding 410 and the                    secondary winding 420.                                                 Rg1 541 Resistance between the primary winding 410 input terminals. The                resistance is due to the conductivity of the dielectric material               used as                                                                        the substrate 360.                                                     Rg2 507 Resistance of dielectric material between the primary winding                  410                                                                            and the secondary winding 420.                                         Rg3 542 Resistance between the secondary winding 420 output terminals.                 The                                                                            resistance is due to the conductivity of the dielectric material               used as                                                                        the substrate 360.                                                     R1oss 509                                                                              Resistance of the ferrite core 450. This is due to a heating of                the core                                                                       when transferring a signal from the primary winding 410 to the                 secondary winding 420.                                                 __________________________________________________________________________

The following describes how the devices, of the integrated transformer linear model 500, are connected. Rg1 540 connects across the input terminals of the integrated transformer. Rs1 510 connects in serial to a first input terminal. L11 520 connects to the other terminal of Rs1 510. L12 525 connects to the other terminal of L11 520. Rs2 515 connects to the other terminal of L12 525. Rg3 542, Lm2 552, Cp2 575, Z2 501 and the primary winding of the ideal transformer 503 connect in parallel between the other terminal of Rs2 515 and the second input terminal. Lm1 550, Cp1 570 and Rloss 509 connect in parallel from the second input terminal and the connection between L11 520 and L12 525. Cc 505 and Rg2 507 connect in parallel to the connection between Rs1 510 and L11 520 and the connection between L12 525 and Rs2 515. The secondary winding of the ideal transformer 503 connects to the output terminals.

In previous systems, where the transformer is used only for isolation, such as in the isolation and filter circuit 100, the leakage inductance is kept to a minimum. That is, prior art transformers are manufactured to reduce the leakage inductance. The reason for this is that the leakage inductor acts to prevent the transformer from transferring all of the input signal to the output. However, in one embodiment of the invention, this feature of the leakage inductance, in conjunction with the primary winding's inter-trace capacitance, are advantageously used to filter the received signal. In this embodiment, the leakage inductance, L11 520, and the inter-trace capacitance, Cp1 570, are a series resonant circuit that act as a low pass filter. To change the resonance frequency, the leakage inductance, L11 520, is varied by varying the distance between the internal edges of the core and the number of turns (greater distance means after leakage inductance). Varying the widths of the primary winding's traces and the distances between these traces varies the inter-trace capacitance, Cp1 570.

Additionally, the magnetizing inductance, Lm1 550, and the inter-trace capacitance, Cp1 570, form a parallel resonant circuit that acts as a band pass filter. By varying Lm1 550 and Cp1 570, the resonance frequency for this filter can be changed. Changing the materials used to make the ferrite core 450, the shape of the core (e.g., circular, rectangular), the layout of the trace, and the number of turns of the trace can vary Lm1 550.

In other embodiments of the invention, Cp1 570 is changed in the following ways. In one embodiment, the substrate 360 includes additional layers, such as a ground plane, that can be used to alter Cp1 570. In another embodiment, extending trace material from the top trace 412 and/or the bottom trace 414 changes Cp1 570. An example of an extended trace is shown in FIG. 5b as extended trace 588. In another embodiment, similar extensions are made on both the primary winding 410 and the secondary winding 420 to vary Cc 505 and Rg2 507.

In one embodiment of the invention, the resonant frequency of both of these resonant circuits is altered so that the integrated transformer passes desirable frequencies and attenuates the other frequencies. Table 2 shows how some of the manufacturing variables can be changed to alter the values of the integrated transformer.

                                      TABLE 2                                      __________________________________________________________________________     Label Dependencies                                                             __________________________________________________________________________     Rs1 510                                                                              Directly proportional to the length of the primary winding 410.                Each                                                                           additional length of the trace material adds to the resistance of              the                                                                            primary winding. Inversely proportional to the width and depth of              the primary winding 410. Also depends on the resistance of the                 trace                                                                          material used.                                                           L11 520                                                                              Depends upon the distance between the internal edges of the core,              the physical layout of the trace, and the number of turns.               Lm1 550                                                                              Directly proportional to the number of turns in the primary                    winding                                                                        410. Depends upon the properties of the ferrite core 450, such as              its                                                                            shape and composition (i.e. composition of metal). Also depends                upon the frequency of input signal, the layout of the trace, and               the                                                                            number of turns.                                                         Lm2 552                                                                              Directly proportional to the number of turns in the secondary                  winding 420. Depends upon the properties of the ferrite core 450,              such as its shape and composition (i.e. composition of metal).                 Also                                                                           depends upon the frequency of input signal, the layout of the                  trace,                                                                         and the number of turns.                                                 Cp1 570                                                                              Inversely proportional to the distance between the traces of the               primary winding 410. This includes the distance between two                    adjacent lengths of the top trace 412 and the distance between the             top trace 412 and the bottom trace 414. In one embodiment, the top             trace is displaced from the bottom trace 414 to alter Cp1 570.                 Also                                                                           depends upon the trace material used and the dielectric                        characteristics of the substrate 360 material.                           Rs2 515                                                                              Similar to Rs1 510 except for the secondary winding 420.                 L12 525                                                                              Similar to L11 520 except for the secondary winding 420.                 Cp2 575                                                                              Similar to Cp1 570 except for the secondary winding 420.                 Cc 505                                                                               Inversely proportional to the distance between the edges of the                primary winding 410 and the secondary winding 420. Depends on                  the trace material used and the dielectric characteristics of the              substrate 360 material. Depends also on the relative trace                     positioning                                                                    of the primary winding 410 and the secondary winding 420. The                  windings are shown adjacent to each other in FIG. 4, however, in               one embodiment, forming the traces of the two windings next to                 each other forms a bifilar transformer.                                  Rg1 540                                                                              Directly proportional to the distance between the edges of the                 primary winding 410 traces. Also depends upon the resistance of                the                                                                            dielectric material used as part of the substrate 360.                   Rg2 507                                                                              Directly proportional to the distance between the edges of the                 primary winding 410 and secondary winding 420. Also depends                    upon the resistance of the dielectric material.                          Rg3 542                                                                              Directly proportional to the distance between the edges of the                 secondary winding 420 traces. Also depends upon the resistance of              the dielectric material used as part of the substrate 360.               R1oss 509                                                                            Depends upon the core material being used as well as the input                 signal excitation (e.g., frequency and voltage).                         __________________________________________________________________________

In one embodiment of the invention, the following manufacturing characteristics are modified: trace width, trace spacing, trace length, winding shape (only in the same approximate planes as the surfaces of the substrate 360), core shape, and core material. Importantly, the characteristics of the trace material and the substrate 360 material are not altered in this embodiment so that the integrated transformer is made using the same trace material and substrate 360 material as is used by, for example, the other adapter electronics 330. That is, in this embodiment, there is a common set of design rules, for making a printed circuit board, that are used by the integrated transformers and the other adapter electronics 330. These design rule specify which parameters may not be changed (e.g., the substrate 360 material, trace thickness) and which parameters may be changed (e.g., trace width).

In other embodiments, reactive elements, such as external capacitors and inductors, are added to the integrated transformer to help achieve a particular frequency response. For example, in one embodiment, an external capacitor, C 535, is included. C 535 is connected across the input terminals to the integrated transformer. For example, in one embodiment, C 535 is mounted through the via 417 and a new via 517. In another embodiment, a surface mounted capacitor is used.

C 535 effectively increases Cp1 570. C 535 is helpful where Cp1 570 cannot be increased to a desired level without undesirably altering other parameters of the model 500. In one embodiment, C 535 is determined so that the band pass filter of Cp1 570, C 535 and Lm1 550 has a frequency response that resonates at 10 MHz. In other embodiments, C 535 values are chosen for other frequencies (e.g., 200 kHz, 4 MHz, 16 MHz, 20 MHz, 100 MHz).

In one embodiment, C 535 is determined so that the integrated transformer, as laid out in FIG. 4, has a resonant frequency of 10 MHz. C 535 is determined as follows.

First, the inductance, the integrated transformer, as measured across the input terminals of the primary winding 410, is determined. L is proportional to L11 520 and Lm1 550. In one embodiment, L is measured, for a given integrated transformer, from a Smith chart plot, at the 10 MHz point, generated by a 1 GHz Hewlett-Packard Network Analyzer. In the example of FIG. 4, L is approximately 643 nH. C is then determined as follows:

ω² =1/LC

C=1/Lω²

C=1/(643 nH)(2 * 3.14 * 10 MHz)²

Thus, C 535 is approximately 395 pF. Thus, this integrated transformer will have a resonant frequency at 10 MHz. (This is shown in FIG. 7a and 7b and described below.)

Integrated Transformer Frequency Response

FIG. 6 illustrates the frequency response of an integrated transformer using the layout of FIG. 4. As can be seen in the 1 GHz integrated transformer frequency response graph 600, at approximately 10 MHz, the signal has been reduced by 1.2 dB. The signal response then begins to roll off, reaching almost -30 dB by approximately 300 MHz. Although not ideal for some communications applications, the graph 600 illustrates that the parasitic characteristics of the integrated transformer can be used to make the integrated transformer into a filter. By altering the parasitic characteristics greater roll off can be achieved and at the desired frequency.

A different Cc 505 changes frequency response. In one embodiment, Cc 505 is reduced to help attenuate undesirable high frequency signals. In one embodiment, Cc 505 is reduced by placing a trace connect to ground between the primary winding 410 and the secondary winding 420. In this embodiment, the additional trace to ground also helps increase Cp1 570 and Cp2 575. In another embodiment, Cc 505 is reduced by increasing the distance between the closest trace edge of the primary winding 410 to the closest trace edge of the secondary winding 420. One method of increasing Cc 505 is to position the primary winding 410 and the secondary winding 420 next to each other to create a bifilar integrated transformer. Cc 505 acts as a parallel resonant circuit with L11 520+L12 525, thereby reducing the energy transfer from the input terminals to the output terminals at a certain frequency.

Integrated Transformer with External Capacitor Frequency Response

FIG. 7a and 7b illustrate the frequency response of an integrated transformer using the layout of FIG. 4 and including an external capacitor C 535 of approximately 380 pF. As can be seen in the 1 GHz integrated transformer frequency response graph 700, at approximately 10 MHz, the signal has been reduce by only 1.3 dB. The signal now begins to roll off much more quickly than the integrated transformer without the capacitor, reaching -20 dB by 40 MHz. Also, advantageously, the high frequency response, at approximately 800 MHz, is also reduced.

FIG. 7b illustrates that only a small amount of the signal is lost in the 100 kHz to 10 MHz frequency range. For ten Mbit Ethernet communications, low signal loss in this frequency range is desirable. However, in one embodiment of the invention, additional changes in the parasitic characteristics of the integrated transformer cause the band stop filter characteristics of L11 and Cp1 to have greater effect. This reduces some of the low frequency components of the signal. For some communications protocols, the loss of these frequency components are relatively insignificant.

FIG. 8a and FIG. 8b illustrate the frequency response of the same embodiment of the integrated transformer as in FIG. 7a and 7b, but with the addition of a second external capacitor C2 545. C2 545 is connected across the output terminals of the integrated transformer to be tuned to the desired frequency response. In one embodiment, the capacitance of C2 545 is determined in a similar manner as C 535. In the example of FIG. 8a and FIG. 8b, C2 545 has a capacitance of approximately 380 pF.

The addition of C2 545, at the secondary winding 420 output terminals, creates a series resonant circuit with L12 525 at approximately 700 MHz. Thus, high frequency signals are attenuated. FIG. 8a shows that all of the high frequency signals are attenuated below 20 dB.

Also note that this embodiment also has significantly better roll off than either the circuits of FIG. 6 or FIG. 7a. FIG. 8b shows that the important frequency response between five MHz and ten MHz is relatively linear. Additionally, the amount of signal loss has improved to only approximately 0.6 dB at approximately 10 MHz. In particular, compared with the prior art frequency response in graph 220, the frequency response of the embodiment of FIG. 8a and FIG. 8b has less signal loss between five MHz and 10 MHz.

FIG. 9 illustrates a bimar transformer as can be used in one embodiment of the invention. In this embodiment, the transformer 910 includes a substrate 920, a core 930, a metal trace 940, and a metal trace 950. The metal trace 940 and the metal trace 950 are formed in a spiral around the core 930. This shape increases the coupling capacitance between the two metal traces.

FIG. 10 illustrates one embodiment of the invention having a transformer 1010 for electrically isolating and filtering a network communications signal. The transformer 1010 comprises a substrate 1020, a closed loop of ferrite core 1030, a metal trace 1040, and a metal trace 1050. The substrate has a hole 1022 and a hole 1024 through the substrate. The closed loop of ferrite core 1030 is disposed through the hole 1022 and the hole 1024. The metal trace 1040 is positioned in a spiral around the hole 1022. The metal trace 1040 is coupled to receive network communications signals. The metal trace 1050 is positioned in a spiral around the hole 1024 and is coupled to filter and transmit network communications signals.

In this embodiment, the substrate has three layers. The metal trace 1040 couples the part of the trace on the top layer to the part of the trace on the middle layer. The part of the trace on the middle layer is coupled to the part of the trace on the bottom layer.

The metal trace 1040 is positioned to at least increase the coupling capacitance to cause the transformer to attenuate signals above 300 MHz by more than three dB while not attenuating signals at, or near 100 MHz, by more than three dB.

Additional Embodiments

FIG. 11 illustrates the traces for an integrated filter transformer. This integrated filter transformer can be used in conjunction with the circuit of FIG. 12 to provide improved filter characteristics for network communications.

FIG. 11 includes four traces. The first pair of traces form the primary winding 1130 of the integrated filter transformer. The second pair of traces form the second winding 1140. The top pair of traces, top traces 1110, are the shape of the traces as they would appear on the top of a printed circuit board. Similarly, the bottom traces 1120 show the traces as they would appear on the bottom of a printed circuit board. In this embodiment the distance between the innermost trace and the edge of each hole is approximately 0.2 inches. In some embodiments, the trace width is 0.007 inches wide. In some embodiments the trace width is 0.008 inches wide with spacing of 0.008 inches.

The primary winding 1130 is coupled to receive the signals transmit plus 1112, center tap 1114 and transmit minus 1116. The secondary winding 1140 is coupled to transmit the cable transmit plus 1160 and the cable transmit minus 1162.

FIG. 12 illustrates one embodiment of an external circuit that can be used with the integrated filter transformer of FIG. 11. In one embodiment of the invention the circuit of FIG. 12 receives signals from an Ethernet controller supplied by 3Com Corporation. In one embodiment such an Ethernet controller includes the Ethernet controller used on the Etherlink 3 (3C509B) family of adapters. This Ethernet controller provides a ten megabit parallel tasking Ethernet communications function.

The core is placed through the holes in the printed circuit board of the integrated transformer. Such cores can be obtained from Tokin, Inc. In some embodiments, these cores include a Tokin FEI 12.5 core that has been cut to remove one of the legs of the core. (The FEI 12.5 core has three legs, as only two legs are needed, the middle leg is removed.)

FIG. 12 includes the following elements: a resistor 1210, a resistor 1220, a capacitor 1230, a capacitor 1240, a resistor 1250, a resistor 1260, a capacitor 1270, a capacitor 1280, a capacitor 1290, a capacitor 1295, a capacitor 1292, a capacitor 1297, a resistor 1212, a resistor 1222, a capacitor 1232, a capacitor 1242, a resistor 1252, a resistor 1262, a capacitor 1272, and a capacitor 1282. These elements are coupled as follows.

A twisted pair pre-emphasis minus 1202 signal is provided to one terminal of the capacitor 1230 and one terminal of the resistor 1210. The other terminal of the capacitor 1230 is coupled to ground. The other terminal of the resistor 1210 is coupled to one terminal of the resistor 1220 and one terminal of the capacitor 1240. The other terminal of the capacitor 1240 is coupled to ground. The other terminal of the resistor 1220 is coupled to one terminal of the capacitor 1290 and one terminal of the capacitor 1295. The twisted pair data plus 104 is coupled to one terminal of the capacitor 1270 and one terminal of the resistor 1250. The other terminal of the capacitor 1270 is coupled to ground. The other terminal of the resistor 1250 is coupled to one terminal of the resistor 1260 and one terminal of the capacitor 1280. The other terminal of the capacitor 1280 is coupled to ground. The other terminal of the resistor 1260 is also coupled to the same terminals of the capacitor 1290 and the capacitor 1295 as resistor 1220 is coupled. The resistor the resistor 1222, the resistor 1252, the resistor 1262, the capacitor 1232, the capacitor 1242, the capacitor 1272, and the capacitor 1282, are similarly configured. The other terminal of the capacitor 1295 is coupled to the terminal of the resistor 1222, the terminal of the resistor 1262 and a terminal of the capacitor 1292. The other terminal of the capacitor 1290 is coupled to a first terminal of the primary winding of the integrated transformer 1200. The other terminal of the capacitor 1292 is coupled to the other terminal of the primary winding of the integrated transformer 1200. The center tap of the integrated transformer 1200 is coupled to ground. The secondary winding terminals are coupled to the terminals of the capacitor 1297 and also to the cable transmit plus 1160 and cable transmit minus 1162 signals.

The integrated transformer 1200 includes parasitic characteristics which filter the signals receive from the Ethernet controller. Additionally, in this embodiment, the circuitry coupled to the primary winding and the capacitor 1297 coupled to the secondary winding provides additional filtering capabilities.

FIG. 13 illustrates the reflected signal transmission for the circuit of FIG. 12. Low reflection values indicate high transmission. The previous response signal 1300 represents a frequency response of the reflected signal from previous systems that do not use such an integrated filter transformer. The new response 1310 shows the reflected signal transmission of the circuit of FIG. 12. As can be seen by the two responses, the new response 1310 has a significantly lower value at the critical frequencies than the old response 1300. This relationship shows that this embodiment has an improved transmission of the important frequencies for ten megabit Ethernet transmissions over prior systems.

FIG. 14 and FIG. 15 illustrate the top and bottom layers of traces for an integrated filter transformer. This transformer has no center tap. Such a transformer could be used on the receive side of the network adapter. In some embodiments of the invention, where the integrated filter transformer of FIG. 14 and FIG. 15 is used, only two and one half windings per trace side are used. In some embodiments, only 2 windings are used on the receive side.

FIG. 16 and FIG. 17 illustrate a center tapped integrated filter transformer as may be used in one embodiment of the invention.

FIG. 18 and FIG. 19 illustrate an integrated filter transformer having a 2:1 ratio and a center tap on the primary winding and no center tap on the secondary winding.

FIG. 20 and FIG. 21 illustrate an integrated circuit transformer having windings spaced relatively close together between the two through holes and relatively far apart on the opposite sides of each winding.

FIG. 22 and FIG. 23 illustrate an alternative configuration with one of the windings on one side of the board and other winding on the other side (no center tap).

FIG. 24 and FIG. 25 illustrate an integrated filter transformer having a center tap where one of the windings is on one side of the board and the other winding is on the other side of the board.

FIG. 26 and FIG. 27 illustrate an integrated filter transformer having one winding on one side of the board around one core and the other winding on the other side of the board around the other side of the core.

FIG. 28 and FIG. 29 illustrate an integrated filter transformer with both primary and secondary windings on the same leg of the ferrite core, but the primary is on top layer of PCB and secondary on bottom layer of PCB.

FIG. 30 and FIG. 31 show an integrated filter transformer having significantly thicker traces than those shown in previous embodiments.

FIG. 32 and FIG. 33 illustrate an alternate way of accessing the center tap as compared to FIG. 16 and 17.

The following describes the width of traces and the spacing between adjacent traces for a number of integrated transformers. These are only examples and the inventions described herein are not limited to only these spacings.

The transformer of FIG. 14 and 15 use standard traces of 0.008 inches wide and 0.008 inches in spacing. The distance between the innermost trace and the corresponding hole is approximately 0.012 inches. In some embodiments, the above spacing requirements will not allow for the number of windings shown in FIG. 14 and FIG. 15 and so the number of turns is reduced by one for each hole. Such spacings apply to the transformers of FIGS. 16 and 17, 18 and 19, 20 and 21, and 26 and 27. The widths of the traces apply to the transformers shown in FIG. 14 through FIG. 27.

In some embodiments of the transformer of FIGS. 26 and 27, the traces are 0.005 inches in width, while the spacing between the traces is kept the same.

The spacing for FIG. 28 and 29 are the same, using 0.008 inches wide traces and 0.008 inches spacing.

The spacing for FIG. 30 and FIG. 31 is 0.01 inches and the traces are 0.012 inches wide.

For FIG. 32 and FIG. 33, the traces are 0.005 inches wide and have a spacing of 0.007 inches.

The above describes many different integrated transformers tuned to provide specific desired frequency responses. Importantly, the integrated transformers greatly reduce the cost of manufacturing filtering and isolation circuits in network adapter cards, and other communications systems, by integrating the filtering and isolation functions into one structure. The integrated transformer is made from traces on a substrate and a ferrite core. In one embodiment, the same techniques used for forming the traces to connect the rest of the components on the substrate are used to create the traces for the windings of the integrated transformer, thereby reducing the cost of the integrated transformer. The parasitic characteristics of the windings and the ferrite core are modified to advantageously change the frequency response of the integrated transformer. In one embodiment, external reactive elements, such as an external capacitor, are added to further change the frequency response of the integrated transformer. 

What is claimed is:
 1. A network adapter card comprising:a substrate having two pairs of holes through said substrate, said substrate having a pattern of a plurality of conductive traces used to form circuits; a network communications circuit, having an input port and an output port, being at least partially formed by at least one trace of said plurality of conductive traces; a receive transformer coupled to said input port; a send transformer coupled to said output port; wherein each of said receive transformer and said send transformer includes,a primary winding, formed from at least a first trace of said plurality of conductive traces, said first trace being formed around at least a first hole of one pair of holes, a secondary winding, formed from at least a second trace of said plurality of conductive traces, said second trace being formed around at least a second hole of said one pair of holes, a core formed in a loop and disposed through one pair of holes of said two pairs of holes, said receive transformer being disposed through a different pair of holes than said send transformer, wherein said primary winding and said secondary winding are formed around a first pair of holes of said pair of holes and said primary winding and said secondary winding are formed around a second pair of holes of said pair of holes, and wherein said first trace and said second trace are positioned to increase a parasitic capacitance of each transformer to cause said send transformer to filter network communication signals received by said primary winding of said send transformer; and wherein said send transformer is coupled to said output port via a filter circuit, said filter circuit for further filtering signals received from said output port prior to said primary winding of said send transformer receiving said signals.
 2. The network adapter card of claim 1 wherein said first trace and said second trace of said send transformer are formed to have a different parasitic capacitance than said first trace and said second trace of said receive transformer, said send transformer having a different frequency response than said receive transformer.
 3. The network adapter card of claim 1 wherein said first trace of said send transformer couples to said output port and wherein said second trace of said receive transformer couples to said input port.
 4. The network adapter card of claim 1 wherein each transformer has a frequency response having less than three dB attenuation at ten MHz and greater than three dB attenuation at greater than 20 MHz.
 5. The network adapter card of claim 1 wherein each transformer has a frequency response having less than three dB attenuation at one hundred MHz and greater than three dB attenuation at greater than three hundred MHz.
 6. The network adapter card of claim 1 further including a first capacitor coupled across said primary winding of said send transformer, said first capacitor to further filter signals received by said primary winding.
 7. The network adapter card of claim 6 wherein said first capacitor is chosen so that said send transformer has a frequency response having less than three dB attenuation at ten MHz and greater than three dB attenuation at greater than approximately fourteen MHz.
 8. The network adapter card of claim 6 farther including a second capacitor coupled across said secondary winding of said send transformer, said second capacitor to further filter signals received by said primary winding.
 9. The network adapter card of claim 1 wherein said first trace is formed partially on a top surface of said substrate and partially on a bottom surface said substrate.
 10. A network adapter card comprising:a circuit board having a first surface and a second surface, said first surface being parallel to said second surface, said circuit board having a pattern of metal traces to form circuits; a communications circuit including at least a first integrated circuit mounted on said circuit board, said communications circuit having an input and an output; a receive transformer, said receive transformer having a first primary winding, a first secondary winding and a first ferrite core,said first primary winding having a first pair of primary winding terminals coupled to receive a signal from a network, said first primary winding being formed from a first spiral metal trace at least partially on or near said first surface, said first secondary winding having a first pair of secondary winding terminals coupled to said output of said communications circuit, said first secondary winding being formed from a second spiral metal trace at least partially on or near said first surface, said first ferrite core being formed in a closed loop through said circuit board and through approximately the centers of said first spiral metal trace and said second spiral metal trace; a send transformer, said send transformer having a second primary winding, a second secondary winding and a second ferrite core,said second primary winding having a second pair of primary winding terminals coupled to said input of said communications circuit, said second primary winding being formed from a third spiral metal trace at least partially on or near said first surface, said second secondary winding having a second pair of secondary winding terminals coupled to transmit a signal from said network adapter card, said second secondary winding being formed from a fourth spiral metal trace at least partially on or near said first surface, said second ferrite core being formed in a closed loop through said circuit board and through approximately the centers of said third spiral metal trace and said fourth spiral metal trace; and wherein said send transformer is coupled to a filter circuit, said send transformer and said filter circuit, in combination, to filter network communication signals to conform to network signal standards prior to said network adapter card transmitting said signals.
 11. The network adapter card of claim 10 wherein said communications circuit includes a circuit for preparing and transmitting ten Mbit Ethernet signals.
 12. The network adapter card of claim 10 wherein said first spiral metal trace and said second spiral metal trace are formed partially on said second surface.
 13. The network adapter card of claim 12 wherein said first spiral metal trace is formed from a first partial metal trace and a second partial metal trace, said first partial metal trace being formed on said first surface, said second partial metal trace being formed on said second surface, and wherein said first partial metal trace is partially offset from said second partial metal trace in a plane parallel with said first surface and said second surface.
 14. The network adapter card of claim 10 wherein said circuit board includes glass epoxy, laminated NEMA GRADE FR-4 material.
 15. The network adapter card of claim 10 wherein said first spiral metal trace, said second spiral metal trace, said third spiral metal trace, are all made of copper, and wherein said first pair of secondary winding terminals are coupled to said input of said communications circuit by a first copper trace, and where said second pair of primary winding terminals are coupled to said output of said communications circuit by a second copper trace. 